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LT1370 500kHz High Efficiency 6A Switching Regulator
FEATURES
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DESCRIPTION
The LT (R)1370 is a monolithic high frequency current mode switching regulator. It can be operated in all standard switching configurations including boost, buck, flyback, forward, inverting and "Cuk." A 6A high efficiency switch is included on the die, along with all oscillator, control and protection circuitry. The LT1370 typically consumes only 4.5mA quiescent current and has higher efficiency than previous parts. High frequency switching allows for very small inductors to be used. New design techniques increase flexibility and maintain ease of use. Switching is easily synchronized to an external logic level source. A logic low on the Shutdown pin reduces supply current to 12A. Unique error amplifier circuitry can regulate positive or negative output voltage while maintaining simple frequency compensation techniques. Nonlinear error amplifier transconductance reduces output overshoot on start-up or overload recovery. Oscillator frequency shifting protects external components during overload conditions.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Faster Switching with Increased Efficiency Uses Small Inductors: 4.7H All Surface Mount Components Low Minimum Supply Voltage: 2.7V Quiescent Current: 4.5mA Typ Current Limited Power Switch: 6A Regulates Positive or Negative Outputs Shutdown Supply Current: 12A Typ Easy External Synchronization Switch Resistance: 0.065 Typ
APPLICATIONS
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Boost Regulators Laptop Computer Supplies Multiple Output Flyback Supplies Inverting Supplies
TYPICAL APPLICATION
5V to 12V Boost Converter
5V L1* D1 MBRD835L
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VOUT 12V
EFFICIENCY (%)
90 88 86 84 82 80 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 LOAD CURRENT (A)
LT1370 * TA02
VIN OFF ON S/S LT1370 VSW
R1 53.6k 1%
+
C1** 22F 25V
FB GND C2 0.047F R3 2k VC
+
R2 6.19k 1% C3 0.0047F
LT1370 * TA01
C4** 22F 25V x2
*COILTRONICS UP2-4R7 (4.7H) UP4-100 (10H) **AVX TPSD226M025R0200 MAX IOUT L1 IOUT 4.7H 1.8A 10H 2A
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12V Output Efficiency
VIN = 5V L = 10H
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LT1370 ABSOLUTE AXI U RATI GS
Operating Ambient Temperature Range ...... 0C to 70C Operating Junction Temperature Range Commercial .......................................... 0C to 125C Industrial ......................................... - 40C to 125C Storage Temperature Range ................ - 65C to 150C Lead Temperature (Soldering, 10 sec)................. 300C Supply Voltage ....................................................... 30V Switch Voltage LT1370 ............................................................... 35V LT1370HV .......................................................... 42V S/S, SHDN, SYNC Pin Voltage ................................ 30V Feedback Pin Voltage (Transient, 10ms) .............. 10V Feedback Pin Current ........................................... 10mA Negative Feedback Pin Voltage (Transient, 10ms) ............................................. 10V
PACKAGE/ORDER I FOR ATIO
FRONT VIEW 7 6 5 4 3 2 1 R PACKAGE 7-LEAD PLASTIC DD VIN NFB VSW GND S/S FB VC TAB IS GND
ORDER PART NUMBER LT1370CR LT1370HVCR LT1370IR LT1370HVIR
TAB IS GND
TJMAX = 125C, JA = 30C/W, JC = 4C/W WITH PACKAGE SOLDERED TO 0.5 INCH2 COPPER AREA OVER BACKSIDE GROUND PLANE OR INTERNAL POWER PLANE. JA CAN VARY FROM 20C/W TO > 40C/W DEPENDING ON MOUNTING TECHNIQUE
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, TA = 25C unless otherwise noted.
SYMBOL VREF IFB PARAMETER Reference Voltage Feedback Input Current Reference Voltage Line Regulation VNFR INFB Negative Feedback Reference Voltage Negative Feedback Input Current Negative Feedback Reference Voltage Line Regulation gm Error Amplifier Transconductance Error Amplifier Source Current Error Amplifier Sink Current CONDITIONS Measured at Feedback Pin VC = 0.8V VFB = VREF
q q
2.7V VIN 25V, VC = 0.8V Measured at Negative Feedback Pin Feedback Pin Open, VC = 0.8V VNFB = VNFR 2.7V VIN 25V, VC = 0.8V IC = 25A
VFB = VREF - 150mV, VC = 1.5V VFB = VREF + 150mV, VC = 1.5V
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FRONT VIEW 7 6 5 4 3 2 1 T7 PACKAGE 7-LEAD TO-220 VIN NFB VSW GND S/S FB VC
ORDER PART NUMBER LT1370CT7 LT1370HVCT7 LT1370IT7 LT1370HVIT7
TJMAX = 125C, JA = 50C/W, JC = 4C/W
MIN 1.230 1.225
TYP 1.245 1.245 250
MAX 1.260 1.265 550 900 0.03 - 2.435 - 2.400 - 15 0.05 1900 2300 350 2400
UNITS V V nA nA %/V V V A %/V mho mho A A
q q q q
0.01 - 2.525 - 2.560 - 45 - 2.48 - 2.48 - 30 0.01 1100 700 120 1500 200 1400
q q q
LT1370
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, TA = 25C unless otherwise noted.
SYMBOL PARAMETER Error Amplifier Clamp Voltage AV f Error Amplifier Voltage Gain VC Pin Threshold Switching Frequency Duty Cycle = 0% 2.7V VIN 25V 0C TJ 125C - 40C TJ 0C (I-Grade)
q q
CONDITIONS High Clamp, VFB = 1V Low Clamp, VFB = 1.5V
MIN 1.5 0.2 0.9 460 440 400 85 35 42 40 6
TYP 1.8 0.3 500 1.1 500 500 95 130
MAX 2.30 0.52 1.35 550 580 580 300
UNITS V V V/ V V kHz kHz kHz % ns V V V
Maximum Switch Duty Cycle Switch Current Limit Blanking Time BV Output Switch Breakdown Voltage
LT1370 q q LT1370HVC, 0C TJ 125C LT1370HVI, - 40C TJ 0C (I-Grade) ISW = 6A Duty Cycle = 50% Duty Cycle = 80% (Note 1)
q q
44 47 0.065 8 7 22 10 0.11 10 33
VSAT ILIM IIN ISW
Output Switch ON Resistance Switch Current Limit Supply Current Increase During Switch ON Time Control Voltage to Switch Current Transconductance Minimum Input Voltage
A A mA/A A/V
q
2.4 4.5 12 0.6 4 -7 600 1.3 12
2.7 6 40 2 25 10 800
V mA A V s A kHz
IQ
Supply Current Shutdown Supply Current Shutdown Threshold Shutdown Delay S/S Input Current Synchronization Frequency Range
2.7V VIN 25V 2.7V VIN 25V, VS/S 0.6V 2.7V VIN 25V 0V S/S 5V
q q q q q q
The q denotes specifications which apply over the full operating temperature range.
Note 1: For duty cycles (DC) between 45% and 85%, minimum switch current limit is given by ILIM = 2.65(2.7 - DC).
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LT1370 TYPICAL PERFORMANCE CHARACTERISTICS
Switch Saturation Voltage vs Switch Current
550 500 450 125C 75C 25C
SWITCH CURRENT LIMIT (A)
SWITCH VOLTAGE (mV)
INPUT VOLTAGE (V)
400 350 300 250 200 150 100 50 0 0 1
0C
4 2 3 SWITCH CURRENT (A)
Shutdown Delay and Threshold vs Temperature
MINIMUM SYNCHRONIZATION VOLTAGE (VP-P)
20 18 16
1.8 SHUTDOWN THRESHOLD 1.6 1.4 1.2 1.0 SHUTDOWN DELAY 0.8 0.6 0.4 0.2 0 0 25 50 75 100 125 150 TEMPERATURE (C)
LT1370 * G04
ERROR AMPLIFIER OUTPUT CURRENT (A)
SHUTDOWN DELAY (s)
14 12 10 8 6 4 2 0 -50 -25
S/S Pin Input Current vs Voltage
2 110
SWITCHING FREQUENCY (% OF TYPICAL)
TRANSCONDUCTANCE (mho)
1 INPUT CURRENT (A) 0 -1 -2 -3 -4 -1 0 1 2 3 4 VOLTAGE (V) 5 6 7
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UW
5 6
LT1370 * G01
Switch Current Limit vs Duty Cycle
8.2 8.0 7.8 7.6 7.4 7.2 7.0 6.8 6.6 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%)
LT1370 * G02
Minimum Input Voltage vs Temperature
3.0 2.8 2.6 2.4 2.2 2.0 1.8 -50 -25
0
25 50 75 100 125 150 TEMPERATURE (C)
LT1370 * G03
Minimum Synchronization Voltage vs Temperature
2.0
Error Amplifier Output Current vs Feedback Pin Voltage
400 300 200 100 0 -100 -200 -300 -0.3 VREF -0.2 -0.1 FEEDBACK PIN VOLTAGE (V) 0.1
LT1370 * G06
3.0 2.5 2.0 1.5 1.0 0.5
fSYNC = 700kHz
25C -55C 125C
SHUTDOWN THRESHOLD (V)
0 -50 -25
0
25 50 75 100 125 150 TEMPERATURE (C)
LT1370 * G05
Switching Frequency vs Feedback Pin Voltage
2000 1800 1600 1400 1200 1000 800 600 400 200
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 FEEDBACK PIN VOLTAGE (V)
LT1370 * G08
Error Amplifier Transconductance vs Temperature
gm = I (VC) V (FB)
100 90 80 70 60 50 40 30 20 10
0 -50 -25
0
25 50 75 100 125 150 TEMPERATURE (C)
LT1370 * G09
LT1370 * G07
LT1370 TYPICAL PERFORMANCE CHARACTERISTICS
VC Pin Threshold and High Clamp Voltage vs Temperature
2.2 2.0
VC VOLTAGE (V)
FEEDBACK INPUT CURRENT (nA)
700 600 500 400 300 200 100
VFB =VREF
NEGATIVE FEEDBACK INPUT CURRENT (A)
1.8 1.6 1.4 1.2
VC HIGH CLAMP
VC THRESHOLD 1.0 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C)
LT1370 * G10
PIN FUNCTIONS
VC: The Compensation pin is used for frequency compensation, current limiting and soft start. It is the output of the error amplifier and the input of the current comparator. Loop frequency compensation can be performed with an RC network connected from the VC pin to ground. See Applications Information. FB: The Feedback pin is used for positive output voltage sensing and oscillator frequency shifting. It is the inverting input to the error amplifier. The noninverting input of this amplifier is internally tied to a 1.245V reference. NFB: The Negative Feedback pin is used for negative output voltage sensing. It is connected to the inverting input of the negative feedback amplifier through a 100k source resistor. S/S: Shutdown and Synchronization Pin. The S/S pin is logic level compatible. Shutdown is active low and the shutdown threshold is typically 1.3V. For normal operation, pull the S/S pin high, tie it to VIN or leave it floating. To synchronize switching, drive the S/S pin between 600kHz and 800kHz. See Applications Information. VIN: Bypass Input Supply Pin with a Low ESR Capacitor, 10F or More. The regulator goes into undervoltage lockout when VIN drops below 2.5V. Undervoltage lockout stops switching and pulls the VC pin low. VSW: The Switch pin is the collector of the power switch and has large currents flowing through it. Keep the traces to the switching components as short as possible to minimize radiation and voltage spikes. GND: Tie all ground pins to a good quality ground plane. See Applications Information.
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Feedback Input Current vs Temperature
800
0
Negative Feedback Input Current vs Temperature
VNFB =VNFR -10
-20
-30
-40
0 -50 -25
0
25 50 75 100 125 150 TEMPERATURE (C)
LT1370 * G11
-50 -50 -25
0
25 50 75 100 125 150 TEMPERATURE (C)
LT1370 * G12
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LT1370
BLOCK DIAGRAM
S/S SYNC OSC LOGIC DRIVER SWITCH
100k NFB 50k
FB
1.245V REF GND SENSE
OPERATION
The LT1370 is a current mode switcher. This means that switch duty cycle is directly controlled by switch current rather than by output voltage. Referring to the block diagram, the switch is turned ON at the start of each oscillator cycle. It is turned OFF when switch current reaches a predetermined level. Control of output voltage is obtained by using the output of a voltage sensing error amplifier to set current trip level. This technique has several advantages. First, it has immediate response to input voltage variations, unlike voltage mode switchers which have notoriously poor line transient response. Second, it reduces the 90 phase shift at midfrequencies in the energy storage inductor. This greatly simplifies closed-loop frequency compensation under widely varying input voltage or output load conditions. Finally, it allows simple pulse-by-pulse current limiting to provide maximum switch protection under output overload or short conditions. A low dropout internal regulator provides a 2.3V supply for all internal circuitry. This low dropout design allows input voltage to vary from 2.7V to 25V with virtually no change in device performance. A 500kHz oscillator is the basic clock for all internal timing. It turns on the output switch via the logic and driver circuitry. Special adaptive antisat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. This minimizes driver dissipation and provides very rapid turn-off of the switch. A 1.245V bandgap reference biases the positive input of the error amplifier. The negative input of the amplifier is brought out for positive output voltage sensing. The error amplifier has nonlinear transconductance to reduce output overshoot on start-up or overload recovery. When the feedback voltage exceeds the reference by 40mV, error amplifier transconductance increases 10 times, which reduces output overshoot. The feedback input also invokes oscillator frequency shifting, which helps protect components during overload conditions. When the feedback voltage drops below 0.6V, the oscillator frequency is reduced 5:1. Lower switching frequency allows full control of switch current limit by reducing minimum switch duty cycle.
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VIN SHUTDOWN DELAY AND RESET LOW DROPOUT 2.3V REG SW ANTI-SAT 5:1 FREQUENCY SHIFT
+
NFBA
-
COMP
+
VC
-
EA AV 20 IA
+
0.005
-
GND
LT1370 * BD
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LT1370
OPERATION
Unique error amplifier circuitry allows the LT1370 to directly regulate negative output voltages. The negative feedback amplifier's 100k source resistor is brought out for negative output voltage sensing. The NFB pin regulates at - 2.48V while the amplifier output internally drives the FB pin to 1.245V. This architecture, which uses the same main error amplifier, prevents duplicating functions and maintains ease of use. Consult LTC Marketing for units that can regulate down to - 1.25V. The error signal developed at the amplifier output is brought out externally. This pin (VC) has three different functions. It is used for frequency compensation, current limit adjustment and soft starting. During normal regulator operation this pin sits at a voltage between 1V (low output current) and 1.9V (high output current). The error amplifier is a current output (gm) type, so this voltage can be externally clamped for lowering current limit. Likewise, a capacitor coupled external clamp will provide soft start. Switch duty cycle goes to zero if the VC pin is pulled below the control pin threshold, placing the LT1370 in an idle mode.
APPLICATIO S I FOR ATIO
Positive Output Voltage Setting
The LT1370 develops a 1.245V reference (VREF) from the FB pin to ground. Output voltage is set by connecting the FB pin to an output resistor divider (Figure 1). The FB pin bias current represents a small error and can usually be ignored for values of R2 up to 7k. The suggested value for R2 is 6.19k. The NFB pin is normally left open for positive output applications. Positive fixed voltage versions are available (consult LTC Marketing). Negative Output Voltage Setting The LT1370 develops a - 2.48V reference (VNFR) from the NFB pin to ground. Output voltage is set by connecting the NFB pin to an output resistor divider (Figure 2). The -30A NFB pin bias current (INFB) can cause output voltage errors and should not be ignored. This has been accounted for in the formula in Figure 2. The suggested value for R2 is 2.49k. The FB pin is normally left open for negative output applications. Dual Polarity Output Voltage Sensing Certain applications benefit from sensing both positive and negative output voltages. One example is the "Dual Output Flyback Converter with Overvoltage Protection" circuit shown in the Typical Applications section. Each output voltage resistor divider is individually set as described above. When both the FB and NFB pins are used,
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VOUT R1 FB PIN R2 VREF
LT1370 * F01
VOUT = VREF 1 + R1 R2 R1 = R2 VOUT -1 1.245
() ()
Figure 1. Positive Output Resistor Divider
-VOUT INFB NFB PIN R2 VNFR R1 = R1
-VOUT = VNFB 1 + R1 + INFB (R1) R2 VOUT- 2.48
()
( )(
2.48 + 30 * 10- 6 R2
)
LT1370 * F02
Figure 2. Negative Output Resistor Divider
the LT1370 acts to prevent either output from going beyond its set output voltage. For example, in this application if the positive output were more heavily loaded than the negative, the negative output would be greater and would regulate at the desired set-point voltage. The positive output would sag slightly below its set-point voltage. This technique prevents either output from going unregulated high at no load.
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LT1370
APPLICATIO S I FOR ATIO
Shutdown and Synchronization
The device has a dual function S/S pin which is used for both shutdown and synchronization. This pin is logic level compatible and can be pulled high, tied to VIN or left floating for normal operation. A logic low on the S/S pin activates shutdown, reducing the part's supply current to 12A. Typical synchronization range is from 1.05 to 1.8 times the part's natural switching frequency, but is only guaranteed between 600kHz and 800kHz. A 12s resetable shutdown delay network guarantees the part will not go into shutdown while receiving a synchronization signal. Caution should be used when synchronizing above 700kHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. This type of subharmonic switching only occurs when the duty cycle of the switch is above 50%. Higher inductor values will tend to eliminate this problem. Thermal Considerations Care should be taken to ensure that the worst-case input voltage and load current conditions do not cause excessive die temperatures. Typical thermal resistance is 30C/W for the R package and 50C/W for the T7 package but these numbers will vary depending on the mounting techniques (copper area, airflow, etc.). Heat is transferred from the package via the tab. Average supply current (including driver current) is: IIN = 4.5mA + DC(ISW/45) ISW = Switch current DC = Switch duty cycle Switch power dissipation is given by: PSW = (ISW)2(RSW)(DC) RSW = Output switch ON resistance Total power dissipation of the die is the sum of supply current times supply voltage, plus switch power: PD(TOTAL) = (IIN)(VIN) + PSW
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Surface mount heat sinks are available which can lower package thermal resistance by two or three times. One manufacturer, Wakefield Engineering, offers surface mount heat sinks for the R package and can be reached at (617) 245-5900 or at www.wakefield.com. Choosing the Inductor For most applications the inductor will fall in the range of 2.2H to 22H. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output current because they reduce peak current seen by the power switch, which has a 6A limit. Higher values also reduce input ripple voltage and reduce core loss. When choosing an inductor you need to consider maximum load current, core and copper losses, allowable component height, output voltage ripple, EMI, fault current in the inductor, saturation and, of course, cost. The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. Assume that the average inductor current for a boost converter is equal to load current times VOUT / VIN and decide whether or not the inductor must withstand continuous overload conditions. If average inductor current at maximum load current is 3A, for instance, a 3A inductor may not survive a continuous 6A overload condition. Also be aware that boost converters are not short-circuit protected and that, under output short conditions, inductor current is limited only by the available current of the input supply. 2. Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, especially with smaller inductors and lighter loads, so don't omit this step. Powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly and other core materials fall in between. The following formula assumes continuous mode operation but it errs only slightly on the high side for discontinuous mode, so it can be used for all conditions.
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LT1370
APPLICATIO S I FOR ATIO
V (V -V ) V IPEAK = (IOUT) OUT + IN OUT IN 2(f)(L)(VOUT) VIN VIN = Minimum input voltage f = 500kHz switching frequency
))
3. Decide if the design can tolerate an "open" core geometry, like a rod or barrel, which has high magnetic field radiation, or whether it needs a closed core, like a toroid, to prevent EMI problems. One would not want an open core next to a magnetic storage media, for instance! This is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. 4. Start shopping for an inductor that meets the requirements of core shape, peak current (to avoid saturation), average current (to limit heating) and fault current. If the inductor gets too hot, wire insulation will melt and cause turn-to-turn shorts. Keep in mind that all good things like high efficiency, low profile and high temperature operation will increase cost, sometimes dramatically. 5. After making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. Use the experts in the LTC Applications Department if you feel uncertain about the final choice. They have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. Output Capacitor The output capacitor is normally chosen by its effective series resistance (ESR), because this is what determines output ripple voltage. At 500kHz any polarized capacitor is essentially resistive. To get low ESR takes volume, so physically smaller capacitors have high ESR. The ESR range needed for typical LT1370 applications is 0.025 to 0.2. A typical output capacitor is an AVX type TPS, 22F at 25V (two each), with a guaranteed ESR less than 0.2. This is a "D" size surface mount solid tantalum capacitor. TPS capacitors are specially constructed and
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tested for low ESR, so they give the lowest ESR for a given volume. To further reduce ESR, multiple output capacitors can be used in parallel. The value in microfarads is not particularly critical, and values from 22F to greater than 500F work well, but you cannot cheat mother nature on ESR. If you find a tiny 22F solid tantalum capacitor, it will have high ESR and output ripple voltage will be terrible. Table 1 shows some typical solid tantalum surface mount capacitors.
Table 1. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current
E CASE SIZE AVX TPS, Sprague 593D AVX TAJ D CASE SIZE AVX TPS, Sprague 593D AVX TAJ C CASE SIZE AVX TPS AVX TAJ B CASE SIZE AVX TAJ 2.5 to 10 0.16 to 0.08 0.2 (Typ) 1.8 to 3.0 0.5 (Typ) 0.22 to 0.17 0.1 to 0.3 0.9 to 2.0 0.7 to 1.1 0.36 to 0.24 ESR (MAX ) 0.1 to 0.3 0.7 to 0.9 RIPPLE CURRENT (A) 0.7 to 1.1 0.4
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Many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents. This is historically true and AVX type TPS capacitors are specially tested for surge capability, but surge ruggedness is not a critical issue with the output capacitor. Solid tantalum capacitors fail during very high turn-on surges, which do not occur at the output of regulators. High discharge surges, such as when the regulator output is dead-shorted, do not harm the capacitors. Single inductor boost regulators have large RMS ripple current in the output capacitor, which must be rated to handle the current. The formula to calculate this is: Output Capacitor Ripple Current (RMS)
DC IRIPPLE (RMS) = IOUT 1 - DC = IOUT VOUT - VIN VIN
DC = Switch duty cycle
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LT1370
APPLICATIO S I FOR ATIO
Input Capacitors
The input capacitor of a boost converter is less critical due to the fact that the input current waveform is triangular and does not contain large squarewave currents as is found in the output capacitor. Capacitors in the range of 10F to 100F with an ESR of 0.1 or less work well up to full 6A switch current. Higher ESR capacitors may be acceptable at low switch currents. Input capacitor ripple current for a boost converter is :
IRIPPLE = 0.3(VIN)(VOUT - VIN) (f)(L)(VOUT)
f = 500kHz switching frequency
The input capacitor can see a very high surge current when a battery or high capacitance source is connected "live" and solid tantalum capacitors can fail under this condition. Several manufacturers have developed tantalum capacitors specially tested for surge capability (AVX TPS series, for instance) but even these units may fail if the input voltage approaches the maximum voltage rating of the capacitor during a high surge. AVX recommends derating capacitor voltage by 2:1 for high surge applications. Ceramic, OS-CON and aluminum electrolytic capacitors may also be used and have a high tolerance to turn-on surges. Ceramic Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low ESR. Unfortunately, the ESR is so low that it can cause loop stability problems. Solid tantalum capacitor ESR generates a loop "zero" at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually resonate with their ESL before ESR becomes effective. They are appropriate for input bypassing because of their high ripple current ratings and tolerance of turn-on surges.
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Output Diode The suggested output diode (D1) is a Motorola MBRD835L. It is rated at 8A average forward current and 35V reverse voltage. Typical forward voltage is 0.4V at 3A. The diode conducts current only during switch OFF time. Peak reverse voltage for boost converters is equal to regulator output voltage. Average forward current in normal operation is equal to output current. Frequency Compensation Loop frequency compensation is performed on the output of the error amplifier (VC pin) with a series RC network. The main pole is formed by the series capacitor and the output impedance (500k) of the error amplifier. The pole falls in the range of 2Hz to 20Hz. The series resistor creates a "zero" at 1kHz to 5kHz, which improves loop stability and transient response. A second capacitor, typically one-tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the VC pin. VC pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. Without the second capacitor, VC pin ripple is:
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VC Pin Ripple =
1.245(VRIPPLE)(gm)(RC) (VOUT)
VRIPPLE = Output ripple (VP-P) gm = Error amplifier transconductance ( 1500mho) RC = Series resistor on VC pin VOUT = DC output voltage
To prevent irregular switching, VC pin ripple should be kept below 50mVP-P. Worst-case VC pin ripple occurs at maximum output load current and will also be increased if poor quality (high ESR) output capacitors are used. The addition of a 0.0047F capacitor on the VC pin reduces switching frequency ripple to only a few millivolts. A low value for RC will also reduce VC pin ripple, but loop phase margin may be inadequate.
LT1370
APPLICATIO S I FOR ATIO
Layout Considerations For maximum efficiency, LT1370 switch rise and fall times are made as short as possible. To prevent radiation and high frequency resonance problems, proper layout of the components connected to the switch node is essential. B field (magnetic) radiation is minimized by keeping output diode, switch pin and output bypass capacitor leads as short as possible. Figure 3 shows recommended positions for these components. E field radiation is kept low by minimizing the length and area of all traces connected to the switch pin. A ground plane should always be used under the switcher circuitry to prevent interplane coupling.
FB VC
GND NFB S/S VSW VIN C C
Figure 3. Layout Considerations-- R Package
L1
SWITCH NODE VOUT
VIN
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The high speed switching current path is shown schematically in Figure 4. Minimum lead length in this path is essential to ensure clean switching and low EMI. The path including the switch, output diode and output capacitor is the only one containing nanosecond rise and fall times. Keep this path as short as possible. More Help For more detailed information on switching regulator circuits, please see Application Note 19. Linear Technology also offers a computer software program, SwitcherCADTM, to assist in designing switching converters. In addition, our Applications Department is always ready to lend a helping hand.
SwitcherCAD is a trademark of Linear Technology Corporation.
D KEEP PATH FROM VSW, OUTPUT DIODE, OUTPUT CAPACITORS AND GROUND RETURN AS SHORT AS POSSIBLE
LT1370 * F03
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HIGH FREQUENCY CIRCULATING PATH
LOAD
LT1370 * F04
Figure 4
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LT1370
TYPICAL APPLICATIONS N
Positive-to-Negative Converter with Direct Feedback
VIN 2.7V TO 13V
OFF
C3 0.0047F
Dual Output Flyback Converter with Overvoltage Protection
R2 6.19k 1% VIN 2.7V TO 10V R1 13k 1% MBRS360T3 T1* 2, 3 7 + P6KE-20A * 1N4148 8, 9 *4 10 * NFB VC GND C2 0.047F R3 2k *DALE LPE-5047-100MB
+
OFF
C3 0.0047F
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T1* 2 D2 P6KE-15A D3 1N4148 1 * 4 *
C1 100F VIN
+
C4 100F x2 R2 2.49k 1% R3 2.49k 1% -VOUT -5V
ON
S/S LT1370
VSW
3
D1 MBRD835L
NFB VC GND C2 0.047F R1 2k
*BH ELECTRONICS 501-0726 MAX I OUT IOUT VIN 1.75A 3V 2.25A 5V 3A 9V
LT1370 * TA03
C1 22F FB ON S/S LT1370 VIN VSW
VOUT 15V C4 47F
+
C5 47F -VOUT -15V R4 12.1k 1% R5 2.49k 1%
1
MBRS360T3
LT1370 * TA04
LT1370
TYPICAL APPLICATIONS N
Two Li-Ion Cells to 5V SEPIC Converter**
VIN 4V TO 9V L1A* 6.8H VIN OFF ON S/S LT1370 VSW
+
C1 33F 20V
+
SINGLE Li-Ion CELL
U
*
C2 4.7F
D1 MBRD835L R2 18.7k 1%
VOUT 5V
FB GND VC
*
L1B* 6.8H
+
R3 6.19k 1%
R1 2k C4 0.047F
C3 100F 10V x2
C5 0.0047F
LT1370 * TA05
C1 = AVX TPSD 336M020R0200 C2 = TOKIN 1E475ZY5U-C304 C3 = AVX TPSD107M010R0100 * BH ELECTRONICS 501-0726 ** INPUT VOLTAGE MAY BE GREATER OR LESS THAN OUTPUT VOLTAGE
MAX I
OUT
IOUT VIN 2A 4V 2.2A 5V 2.6A 7V 2.8A 9V
Single Li-Ion Cell to 5V
D1 MBRD835L VOUT 5V R1 18.7k 1%
L1*
VIN OFF ON S/S LT1370 VSW
+
C1** 100F 10V
FB GND C2 0.047F R3 2k VC
+
R2 6.19k 1% C3 0.0047F
LT1370 * TA06
C4** 100F 10V x2
*COILCRAFT DO3316P-472 **AVX TPSD107M010R0100
MAX I OUT
IOUT 2.5A 3A 3.3A
VIN 2.7V 3.3V 3.6V
13
LT1370
TYPICAL APPLICATIONS N
Laser Power Supply
0.01F 5kV 1800pF 10kV
L1 5 4 1
11 2
Q1
150 MUR405
VIN 12V TO 25V
VSW VIN FB LT1370 2.2F VC GND
+
+
10F
LT1370 * TA07
14
U
47k 5W
1800pF 10kV
8 3
HV DIODES LASER
+
2.2F 0.47F Q2
L2 82H
10k 0.1F
10k VIN 1N4002 (ALL) 190 1% L1 = COILTRONICS CTX02-11128 L2 = GOWANDA GA40-822K Q1, Q2 = ZETEX ZTX849 0.47F = WIMA 3X 0.15F TYPE MKP-20 HV DIODES = SEMTECH-FM-50 LASER = HUGHES 3121H-P COILTRONICS (407) 241-7876
LT1370
PACKAGE DESCRIPTION
0.256 (6.502)
0.060 (1.524)
0.060 (1.524)
0.183 (4.648)
0.075 (1.905) 0.300 (7.620) BOTTOM VIEW OF DD PAK HATCHED AREA IS SOLDER PLATED COPPER HEAT SINK +0.012 0.143 -0.020 0.040 - 0.060 (1.016 - 1.524) 0.026 - 0.036 (0.660 - 0.914) 0.013 - 0.023 (0.330 - 0.584)
0.390 - 0.415 (9.906 - 10.541)
0.460 - 0.500 (11.684 - 12.700)
0.040 - 0.060 (1.016 - 1.524)
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
U
Dimensions in inches (millimeters) unless otherwise noted.
R Package 7-Lead Plastic DD Pak
(LTC DWG # 05-08-1462)
0.060 (1.524) TYP 0.390 - 0.415 (9.906 - 10.541) 15 TYP 0.059 (1.499) TYP
0.165 - 0.180 (4.191 - 4.572)
0.045 - 0.055 (1.143 - 1.397) +0.008 0.004 -0.004
0.330 - 0.370 (8.382 - 9.398)
(
+0.203 0.102 -0.102
)
0.095 - 0.115 (2.413 - 2.921) 0.050 0.012 (1.270 0.305)
(
+0.305 3.632 -0.508
)
R (DD7) 0396
T7 Package 7-Lead Plastic TO-220 (Standard)
(LTC DWG # 05-08-1422)
0.147 - 0.155 (3.734 - 3.937) DIA 0.230 - 0.270 (5.842 - 6.858) 0.570 - 0.620 (14.478 - 15.748) 0.330 - 0.370 (8.382 - 9.398) 0.620 (15.75) TYP 0.700 - 0.728 (17.780 - 18.491) 0.165 - 0.180 (4.191 - 4.572)
0.045 - 0.055 (1.143 - 1.397)
0.152 - 0.202 0.260 - 0.320 (3.860 - 5.130) (6.604 - 8.128) 0.026 - 0.036 (0.660 - 0.914) 0.135 - 0.165 (3.429 - 4.191)
0.095 - 0.115 (2.413 - 2.921)
0.013 - 0.023 (0.330 - 0.584) 0.155 - 0.195 (3.937 - 4.953)
T7 (TO-220) (FORMED) 1197
15
LT1370 RELATED PARTS
PART NUMBER LT1171 LTC 1265 LT1302 LT1372 LT1373 LT1374 LT1376 LT1512 LT1513
(R)
DESCRIPTION 100kHz 2.5A Boost Switching Regulator 12V 1.2A Monolithic Buck Converter Micropower 2A Boost Converter 500kHz 1.5A Boost Switching Regulator Low Supply Current 250kHz 1.5A Boost Switching Regulator 500kHz 4.5A Buck Switching Regulator 500kHz 1.5A Buck Switching Regulator 500kHz 1.5A SEPIC Battery Charger 500kHz 3A SEPIC Battery Charger
COMMENTS Good for Up to VIN = 40V Converts 5V to 3.3V at 1A with 90% Efficiency Converts 2V to 5V at 600mA in SO-8 Packages Also Regulates Negative Flyback Outputs 90% Efficient Boost Converter with Constant Frequency Converts 12V to 3.3V at 2.5A in SO-8 Package Steps Down from Up to 25V Using 4.7H Inductors Input Voltage May Be Greater or Less Than Battery Voltage Input Voltage May Be Greater or Less Than Battery Voltage
16
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417 q (408) 432-1900 FAX: (408) 434-0507q TELEX: 499-3977 q www.linear-tech.com
1370f LT/TP 0198 4K * PRINTED IN THE USA
(c) LINEAR TECHNOLOGY CORPORATION 1998

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